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FEATURES 12-Bit Resolution 24-Pin "Skinny DIP" Package Conversion Time: 500 ns max--AD671J/K/S-500 Conversion Time: 750 ns max--AD671J/K/S-750 Low Power: 475 mW Unipolar (0 V to +5 V, 0 V to +10 V) and Bipolar Input Ranges ( 5 V) Twos Complement or Offset Binary Output Data Out-of-Range Indicator MIL-STD-883 Compliant Versions Available
AIN BPO/UPO 20 21 RANGE SELECT
Monolithic 12-Bit 2 MHz A/D Converter AD671
FUNCTIONAL BLOCK DIAGRAM
ENCODE REF IN VCC ACOM VEE 16 19 23 22 24 V LOGIC 17 DCOM 18
X4 3-BIT FLASH COARSE 4-BIT FLASH 4 8-BIT LADDER MATRIX FINE 4-BIT FLASH 4
3-BIT FLASH
DAC 3
DAC 3
CORRECTION LOGIC 8 LATCHES
AD671
12 14 13 12 1 BIT1-12 15 DAV
OTR MSB
PRODUCT DESCRIPTION
PRODUCT HIGHLIGHTS
The AD671 is a high speed monolithic 12-bit A/D converter offering conversion rates of up to 2 MHz (500 ns conversion time). The combination of a merged high speed bipolar/CMOS process and a novel architecture results in a combination of speed and power consumption far superior to previously available hybrid implementations. Additionally, the greater reliability of monolithic construction offers improved system reliability and lower costs than hybrid designs. The AD671 uses a subranging flash conversion technique, with digital error correction for possible errors introduced in the first part of the conversion cycle. An on-chip timing generator provides strobe pulses for each of the four internal flash cycles and assures adequate settling time for the interflash residue amplifier. A single ENCODE pulse is used to control the converter. The performance of the AD671 is made possible by using high speed, low noise bipolar circuitry in the linear sections and low power CMOS for the logic sections. Analog Devices' ABCMOS-1 process provides both high speed bipolar and 2-micron CMOS devices on a single chip. Laser trimmed thin-film resistors are used to provide accuracy and temperature stability. The AD671 is available in two conversion speeds and performance grades. The AD671J and K grades are specified for operation over the 0C to +70C temperature range. The AD671S grades are specified for operation over the -55C to +125C temperature range. All grades are available in a 0.300 inch wide 24-pin ceramic DIP. The J and K grades are also available in a 24-pin plastic DIP.
1. The AD671 offers a single chip 2 MHz analog-to-digital conversion function in a space saving 24-pin DIP. 2. Input signal ranges are 0 V to +5 V and 0 V to +10 V unipolar, and -5 V to +5 V bipolar, selected by pin strapping. Input resistance is 1.5 k. Power supplies are +5 V and -5 V, and typical power consumption is less than 500 mW. 3. The external +5 V reference can be chosen to suit the dc accuracy and temperature drift requirements of the application. 4. Output data is available in unipolar, bipolar offset or bipolar twos complement binary format. 5. An OUT OF RANGE output bit indicates when the input signal is beyond the AD671's input range. 6. The AD671 is available in versions compliant with the MILSTD-883. Refer to the Analog Devices Military Products Databook or current AD671/883B data sheet for detailed specifications.
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Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
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AD671-SPECIFICATIONS (T to T with V
MIN MAX
DC SPECIFICATIONS unless otherwise noted)
Parameter RESOLUTION ACCURACY (+25C) Integral Nonlinearity (INL) TMIN to TMAX Differential Nonlinearity (DNL) TMIN to TMAX No Missing Codes Unipolar Offsetl Bipolar Zerol Gain Error2 TEMPERATURE COEFFICIENTS3 Unipolar Offset Bipolar Zero Gain Error ANALOG INPUT Input Ranges Bipolar Unipolar Input Resistance 10 Volt Range 5 Volt Range Input Capacitance Reference Input Resistance POWER SUPPLIES Power Supply Rejection4 VCC (+5 V 0.25 V) VLOGIC (+5 V 0.5 V) VEE (-5 V 0.25 V) Operating Voltages VCC VLOGIC VEE Operating Current ICC ILOGIC5 IEE POWER CONSUMPTION TEMPERATURE RANGE Specified (J/K) Specified (S) 0 -55 Min 12
CC
= +5 V
5%, VLOGIC = +5 V
10%, VEE = -5 V
5%, VREF = +5.000 V,
AD671J/S-500 Typ
Max
Min 12
AD671K-500 Typ
Max
Units Bits
4 10 10 Bits Guaranteed 4 10 0.25 10 15 20 11 11 Bits Guaranteed
2
LSB Bits
0.1
0.1
4 10 0.25 10 15 20
LSB LSB % FSR ppm/C ppm/C ppm/C
-5 0 0 1.0 0.5 2.4 1.5 0.75 10 3.5
+5 +5 +10 2.0 1.0 4.7
-5 0 0 1.0 0.5 2.4 1.5 0.75 10 3.5
+5 +5 +10 2.0 1.0 4.7
Volts Volts Volts k k pF k
1 1 1 +4.75 +4.5 -5.25 46 3 46 475 +5.25 +5.5 -4.75 56 6 56 621 +70 +125 0 +4.75 +4.5 -5.25 46 3 46 475
1 1 1 +5.25 +5.5 -4.75 56 6 56 621 +70
LSB LSB LSB Volts Volts Volts mA mA mA mW C C
NOTES 1 Adjustable to zero with external potentiometers. See Offset/Gain Calibration section for additional information. 2 Full-scale range (FSR) is 5 V for the 0 V to 5 V range and 10 V for the 0 V to 10 V and -5 V to +5 V ranges. 3 25C to TMIN and 25C to TMAX. 4 Change in gain error as a function of the dc supply voltage. 5 Tested under static conditions. See Figure 12 for typical curves of ILOGIC vs. Conversion Rate and Output Loading. Specifications subject to change without notice. Specifications shown in boldface are tested on all devices at final electrical test with worst case supply voltages at 0, +25 C and +70C. Results from those tests are used to calculate outgoing quality levels. All min and max specifications are guaranteed, although only those shown in boldface are tested.
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AD671 DC SPECIFICATIONS unless otherwise noted)
Parameter RESOLUTION ACCURACY (+25C) Integral Nonlinearity (INL) TMIN to TMAX (J) TMIN to TMAX (S) Differential Nonlinearity (DNL) TMIN to TMAX No Missing Codes Unipolar Offsetl Bipolar Zerol Gain Error2 TEMPERATURE COEFFICIENTS3 Unipolar Offset Bipolar Zero Gain Error ANALOG INPUT Input Ranges Bipolar Unipolar Input Resistance 10 Volt Range 5 Volt Range Input Capacitance Reference Input Resistance POWER SUPPLIES Power Supply Rejection4 VCC (+5 V 0.25 V) VLOGIC (+5 V 0.5 V) VEE (-5 V 0.25 V) Operating Voltages Vcc VLOGIC VEE Operating Current ICC ILOGIC5 IEE POWER CONSUMPTION TEMPERATURE RANGE Specified (J/K) Specified (S) 0 -55 Min 12
(TMIN to TMAX with VCC = +5 V
5%, VLOGIC = +5 V
10%, VEE = -5 V
5%, VREF = +5.000 V,
AD671J/S-750 Typ
Max
Min 12
AD671K-750 Typ
Max
Units Bits
2 2.5 11 11 Bits Guaranteed 4 10 0.25 10 15 20 12 12 Bits Guaranteed
1.5
LSB LSB Bits
0.1
0.1
4 10 0.25 10 15 20
LSB LSB % FSR ppm/C ppm/C ppm/C
-5 0 0 1.0 0.5 2.4 1.5 0.75 10 3.5
+5 +5 +10 2.0 1.0 4.7
-5 0 0 1.0 0.5 2.4 1.5 0.75 10 3.5
+5 +5 +10 2.0 1.0 4.7
Volts Volts Volts k k pF k
1 1 1 +4.75 +4.5 -5.25 46 3 46 475 +5.25 +5.5 -4.75 56 6 56 621 +70 +125 0 +4.75 +4.5 -5.25 46 3 46 475
1 1 1 +5.25 +5.5 -4.75 56 6 56 621 +70
LSB LSB LSB Volts Volts Volts mA mA mA mW C C
NOTES 1 Adjustable to zero with external potentiometers. See Offset/Gain Calibration section for additional information. 2 Full-scale range (FSR) is 5 V for the 0 V to 5 V range and 10 V for the 0 V to 10 V and -5 V to +5 V ranges. 3 25C to TMIN and 25C to TMAX. 4 Change in gain error as a function of the dc supply voltage. 5 Tested under static conditions. See Figure 12 for typical curves of I LOGIC vs. Conversion Rate and Output Loading. Specifications subject to change without notice. Specifications shown in boldface are tested on all devices at final electrical test with worst case supply voltages at 0, +25 C and +70C. Results from those tests are used to calculate outgoing quality levels. All min and max specifications are guaranteed, although only those shown in boldface are tested.
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AD671-SPECIFICATIONS T (For all grades
DIGITAL SPECIFICATIONS
Parameter LOGIC INPUT High Level Input Voltage Low Level Input Voltage High Level Input Current (VIN = VLOGIC) Low Level Input Current (VIN = 0 V) Input Capacitance LOGIC OUTPUTS High Level Output Voltage (IOH = 0.5 mA) Low Level Output Voltage (IOL = 1.6 mA) Output Capacitance
5%, VLOGIC = +5 V MIN to TMAX, with VCC = +5 V 5%, VREF = +5.000 V, unless otherwise noted)
Symbol VIH VIL IIH IIL CIN VOH VOL COUT Min +2.0 -10 -10 5 +2.4 Typ
10%, VEE = -5 V
Max Units V V A A pF V V pF
+0.8 +10 +10
+0.4 5
Specifications shown in boldface are tested on all devices at final electrical test. Results from those tests are used to calculate outgoing quality levels. All min and max specifications are guaranteed, although only those shown in boldface are tested. Specifications subject to change without notice.
SWITCHING SPECIFICATIONS
Parameter
(For all grades TMIN to TMAX with VCC = +5 V 5%, VLOGIC = +5 V 5%, VIL = 0.8 V, VIH = 2.0 V, VOL = 0.4 V and VOH = 2.4 V)
Symbol tC tC tENC tENC tENCL tDAV tDAV tF tR tDD1 tSS2 20 20 20 75 75 0 0 20 20 Min Typ 475 725
10%, VEE = -5 V
Max 500 750 30 50 Units ns ns ns ns ns ns ns ns ns ns ns
Conversion Time (AD671-500) (AD671-750) ENCODE Pulse Width High (AD671-500) (AD671-750) ENCODE Pulse Width Low DAV Pulse Width (AD671-500) (AD671-750) ENCODE Falling Edge Delay Start New Conversion Delay Data and OTR Delay from DAV Falling Edge Data and OTR Valid before DAV Rising Edge
200 300
75 75
NOTES 1 tDD is measured from when the falling edge of DAV crosses 0.8 V to when the output crosses 0.4 V or 2.4 V with a 25 pF load capacitor on each output pin. 2 tSS is measured from when the outputs cross 0.4 V or 2.4 V to when the rising edge of DAV crosses 2.4 V with a 25 pF load capacitor on each output pin.
a. Encode Pulse HIGH Figure 1. AD671 Timing Diagrams
b. Encode Pulse LOW
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ABSOLUTE MAXIMUM RATINGS* ORDERING GUIDE
Parameter VCC VEE VLOGIC ACOM VCC ENCODE REF IN AIN, BPO/UPO Junction Temperature Storage Temperature Lead Temperature (10 sec) Power Dissipation
With Respect to Min ACOM ACOM DCOM DCOM VLOGIC DCOM ACOM ACOM -0.5 -6.5 -0.5 -1.0 -6.5 -0.5 -0.5 -6.5
Modell Max Units AD671JD-500 AD671KD-500 AD671JD-750 AD671KD-750 AD671SD-500 AD671SD-750
Linearity 4 LSB 2 LSB 2 LSB 1.5 LSB 4 LSB 2.5 LSB
Temperature Range 0C to +70C 0C to +70C 0C to +70C 0C to +70C -55C to +125C -55C to +125C
Package Options2 D-24A D-24A D-24A D-24A D-24A D-24A
+6.5 Volts +0.5 Volts +6.5 Volts +1.0 Volts +6.5 Volts VLOGIC +0.5 Volts VCC +0.5 Volts 11.0 Volts +175 C -65 +150 C +300 C 1000 mW
NOTES 1 For details on grade and package offerings screened in accordance with MIL-STD-883, refer to the Analog Devices Military Products Databook or current AD671/883 data sheet. 2 D = Ceramic DIP.
*Stresses above those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for extended periods may effect device reliability.
CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD671 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
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AD671
AD671 PIN DESCRIPTION
Symbol ACOM AIN BIT1 (MSB) BIT2-BIT11 BIT12 (LSB) BPO/UPO
Pin 22 20 12 11-2 1 21
Type Name and Function P AI DO DO DO AI Analog Ground. Analog Input Signal. Most Significant Bit. Data Bits 2-11. Least Significant Bit. Bipolar or Unipolar Configuration Pin. Connect to AIN for 0 V to +5 V Span, to ACOM for 0 V to +10 V Span and to REF IN for -5 V to +5 V Span. Data Available Output. The Rising Edge of DAV Indicates an End of Conversion and Can Be Used to Latch Current Data into an External Register. The Falling Edge of DAV Can Be Used to Latch Previous Data into an External Register. Digital Ground. The AD671 Starts a Conversion on the Rising Edge of the ENCODE Pulse. Inverted Most Significant Bit. Provides Twos Complement Output Data Format. Out of Range Is Active HIGH when the analog input is beyond the input range of the converter. +5 V Reference Input. +5 V Analog Power. -5 V Analog Power. +5 V Digital Power.
CONNECTION DIAGRAM PINOUT
BIT12 (LSB)
1
24 VEE 23 VCC 22 ACOM 21 BPO/UPO 20 AIN
BIT11 2 BIT10 BIT9 BIT8 BIT7 BIT6 BIT5 BIT4 3 4 5 6 7 8 9
AD671
TOP VIEW (Not to Scale)
19 REF IN 18 DCOM 17 VLOGIC 16 ENCODE 15 DAV 14 OTR 13 MSB
DAV
15
DO
BIT3 10 BIT2 11 BIT1 (MSB) 12
DCOM ENCODE
18 16
P DI
MSB
13
DO
OTR
14
DO
REF IN VCC VEE VLOGIC
19 23 24 17
AI P P P
TYPE: AI = Analog Input DI = Digital Input DO = Digital Output P = Power
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AD671
DEFINITIONS OF SPECIFICATIONS
INTEGRAL NONLINEARITY (INL) GAIN ERROR
Integral nonlinearity refers to the deviation of each individual code from a line drawn from "zero" through "full scale." The point used as "zero" occurs 1/2 LSB (1.22 mV for a 10 V span) before the first code transition (all zeros to only the LSB on). "Full scale" is defined as a level 1 1/2 LSB beyond the last code transition (to all ones). The deviation is measured from the low side transition of each particular code to the true straight line.
DIFFERENTIAL NONLINEARITY (DNL, NO MISSING CODES)
The last transition (from 1111 1111 1110 to 1111 1111 1111) should occur for an analog value 1 1/2 LSB below the nominal full scale (9.9963 volts for 10.000 volts full scale). The gain error is the deviation of the actual level at the last transition from the ideal level. The gain error can be adjusted to zero as shown in Figures 7, 8 and 9.
TEMPERATURE COEFFICIENTS
The temperature coefficients for unipolar offset, bipolar zero and gain error specify the maximum change from the initial (+25C) value to the value at TMIN or TMAX.
POWER SUPPLY REJECTION
An ideal ADC exhibits code transitions that are exactly 1 LSB apart. DNL is the deviation from this ideal value. Thus every code must have a finite width. Guaranteed no missing codes to 10-bit resolution indicates that all 1024 codes represented by Bits 1-10 must be present over all operating ranges. Guaranteed no missing codes to 11- or 12-bit resolution indicates that all 2048 and 4096 codes, respectively, must be present over all operating ranges.
UNIPOLAR OFFSET
The only effect of power supply error on the performance of the device will be a small change in gain. The specifications show the maximum full-scale change from the initial value with the supplies at the various limits.
SIGNAL-TO-NOISE AND DISTORTION (S/N+D) RATIO
The first transition should occur at a level 1/2 LSB above analog common. Unipolar offset is defined as the deviation of the actual from that point. This offset can be adjusted as discussed later. The unipolar offset temperature coefficient specifies the maximum change of the transition point over temperature, with or without external adjustments.
BIPOLAR ZERO
S/N+D is the ratio of the rms value of the measured input signal to the rms sum of all other spectral components, including harmonics but excluding dc. The value for S/N+D is expressed in decibels.
EFFECTIVE NUMBER OF BITS (ENOB)
ENOB is calculated from the expression SNR = 6.02N + 1.8 dB, where N is equal to the effective number of bits.
TOTAL HARMONIC DISTORTION (THD)
In the bipolar mode the major carry transition (0111 1111 1111 to 1000 0000 0000) should occur for an analog value 1/2 LSB below analog common. The bipolar offset error and temperature coefficient specify the initial deviation and maximum change in the error over temperature.
THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal and is expressed as a percentage or in decibels.
PEAK SPURIOUS OR PEAK HARMONIC COMPONENT
The peak spurious or peak harmonic component is the largest spectral component excluding the input signal and dc. This value is expressed in decibels relative to the rms value of a fullscale input signal.
Theory of Operation
The AD671 uses a successive subranging architecture. The analog to digital conversion takes place in four independent steps or flashes. The analog input signal is subranged to an intermediate residue voltage for the final 12-bit result by utilizing multiple flashes with subtraction DACs (see the AD671 functional block diagram). The AD671 can be configured to operate with unipolar (0 V to +5 V, 0 V to +10 V) or bipolar ( 5 V) inputs by connecting AIN (Pin 20), REFIN (Pin 19) and BPO/UPO (Pin 21) as shown in Figure 2. The AD671 conversion cycle begins by simply providing an active HIGH pulse on the ENCODE pin (Pin 16). The rising edge of the ENCODE pulse starts the conversion. The falling edge of the ENCODE pulse is specified to operate within a window of time: less than 30 ns after the rising edge of ENCODE (AD671-500) and less than 50 ns after the falling edge of ENCODE (AD671-750) or after the falling edge of DAV. The time window prevents digitally coupled noise from being introduced during the final stages of conversion. An internal timing generator circuit accurately controls all internal timing.
ACOM 22 BPO/UPO 21 AIN 20 REF IN 19 BPO/UPO 21 AIN 20 REF IN 19 BPO/UPO 21 AIN 20 REF IN 19
AIN + 5V REF
AIN + 5V REF
AIN
+ 5V REF
0 TO + 5V
0 TO +10V
- 5V TO + 5V
Figure 2. Input Range Connections
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AD671
Upon receipt of an ENCODE command, the first 3-bit flash converts the analog input voltage. The 3-bit result is passed to a correction logic register and a segmented current output DAC. The DAC output is connected through a resistor (within the Range/Span Select Block) to AIN. A residue voltage is created by subtracting the DAC output from AIN, which is less than one eighth of the full-scale analog input. The second flash has an input range that is configured with one bit of overlap with the previous DAC. The overlap allows for errors during the flash conversion. The first residue voltage is connected to the second 3-bit flash and to the noninverting input of a high speed, differential, gain-of-four amplifier. The second flash result is passed to the correction logic register and to the second segmented current output DAC. The output of the second DAC is connected to the inverting input of the differential amplifier. The differential amplifier output is connected to a two step backend 8-bit flash. This 8-bit flash consists of coarse and fine flash converters. The result of the coarse 4-bit flash converter, also configured to overlap one bit of DAC 2, is connected to the correction logic register and selects one of 16 resistors from which the fine 4-bit flash will establish its span voltage. The fine 4-bit flash is connected directly to the output latches. The AD671 will flag an out-of-range condition when the input voltage exceeds the analog input range. OTR (Pin 14) is active HIGH when an out of range high or low condition exists. Bits 1-12 are HIGH when the analog input voltage is greater than the selected input range and LOW when the analog input is less than the selected input range.
INPUT BUFFER AMPLIFIER
The closed-loop output impedance of an op amp is equal to the open loop output impedance (usually a few hundred ohms) divided by the loop gain at the frequency of interest. It is often assumed that loop gain of a follower-connected op amp is sufficiently high to reduce the closed-loop output impedance to a negligibly small value, particularly if the input signal is low frequency. At higher frequencies the open-loop gain is lower, increasing the output impedance which decreases the instantaneous analog input voltage and produces an error. The recommended wideband, fast settling input amplifiers for use with the AD671 are the AD841, AD843, AD845 or the AD847. The AD841 is unity gain stable and recommended as a follower connected op amp. The AD843 and AD845 FET inputs make them ideal for high speed sample-and-hold amplifiers and the AD847 can be used as a low power, high speed buffer. Figure 4 shows the AD841 driving the AD671. As shown in the figure the analog input voltage should be produced with respect to the ACOM pin.
23 4 + 5V - 22 ACOM 18 DCOM + 5V REF 11 VCC 20 AIN
17 24 VEE VLOGIC BIT1 1 BIT12 12 ENCODE 16 DAV 15 OTR 14 MSB 13
AD841 10 5 6
APPLYING THE AD671
DRIVING THE AD671 ANALOG INPUT
19 REF IN 21 BPO/UPO
The AD671 uses a very high speed current output DAC to subtract a known voltage from the analog input. This results in very fast steps of current at the analog input. It is important to recognize that the signal source driving the analog input of the AD671 must be capable of maintaining the input voltage under dynamically-changing load conditions. When the AD671 starts its conversion cycle, the subtraction DAC will sink up to 5 mA (see Figure 3) from the source driving the analog input. The source must respond to this current step by settling the input voltage back to a fraction of an LSB before the AD671 makes its final 12-bit decision.
AD671
Figure 4. Input Buffer Amplifier
REFERENCE INPUT
+ -
IIN
R
A/D IA/D
DAC IDAC
AD671
Figure 3. Driving the Analog Input
Unlike successive approximation A/Ds, where the input voltage must settle to a fraction of a 12-bit LSB before each successive bit decision is made, the AD671 requires the analog input voltage settle to within 12 bits before the third flash conversion, approximately 200 ns. This "free" 200 ns is useful in applications requiring a sample-and-hold amplifier (SHA), overlapping the SHA's hold mode settling time within the 200 ns window will increase total system throughput. See the "Discrete Sampleand-Hold" section for a high speed SHA application.
The AD671 uses a standard +5 volt reference. The initial accuracy and temperature stability of the reference can be selected to meet specific system requirements. Like the analog input, fast switching input-dependent currents are modulated at the reference input pin (REF IN-Pin 19). However, unlike the analog input the reference input is held at a constant +5 volts with the use of capacitor. The recommended reference is the AD586, a +5 V precision reference with an output buffer amplifier. Figure 5 shows the AD671 configured in the 5 V input range. The 6.8 F capacitor maintains a constant +5 volts under the dynamically changing load conditions. An optional 1 F noise reduction capacitor can be connected to the AD586, further reducing broadband output noise. To minimize ground voltage drops the AD586's ground pin should be tied as close as possible to the AD671's ACOM pin. See Figures 20, 21 and 22 for PCB layout recommendations.
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AD671
23 VCC 20 AIN +15V 5V 24 17 VEE VLOGIC BIT1 1 BIT12 12 ENCODE 16 DAV 15 OTR 14 MSB 13
Table I. Grounding and Decoupling Guidelines
Power Supply Decoupling Capacitor Values
Comment 0.1 F (Ceramic) and 10 F (Tantalum). (Surface Mount Chip Capacitors Recommended to Reduce Lead Inductance).
U3
22 ACOM 18 DCOM 19 REF IN 21 BPO/UPO
8 1F C14
VOUT NOISE 6 REDUCTION 6.8F C15 GND 4
AD586 U4
2 +VIN
Capacitor Locations Directly at Positive and Negative Supply Pins to Respective Ground Plane. Grounding Analog Ground Ground Plane or Wide Ground Return Connected to the Analog Power Supply. Ground Plane or Wide Ground Return Connected to the Digital Power Supply. Connected Together Once at the AD671.
AD671
Figure 5. AD586 as Reference Input for AD671
Digital Ground
GROUNDING AND DECOUPLING RULES
Proper grounding and decoupling should be a primary design objective in any high speed, high resolution system. The AD671 separates analog and digital grounds to optimize the management of analog and digital ground currents in a system. The AD671 is designed to minimize the current flowing from ACOM (Pin 22) by directing the majority of the current from VCC (+5 V-Pin 23) to VEE (-5 V-Pin 24). Minimizing analog ground currents hence reduces the potential for large ground voltage drops. This can be especially true in systems that do not utilize ground planes or wide ground runs. ACOM is also configured to be code independent, therefore reducing input dependent analog ground voltage drops and errors. The input current supplied by the external reference (REFIN-Pin 19) and the majority of the full-scale input signal (AIN-Pin 20) are also directed to VEE. Also critical in any high speed digital design are the use of proper digital grounding techniques to avoid potential CMOS "ground bounce." Figure 6 is provided to assist in the proper layout, grounding and decoupling techniques. Table I is a list of grounding and decoupling guidelines that should be reviewed before laying out a printed circuit board.
+ 5V 10F - 5V 10F + 5V 10F
Analog and Digital Ground
UNIPOLAR (0 V TO +10 V) CALIBRATION
The AD671 is factory trimmed to minimize offset, gain and linearity errors. In some applications the offset and gain errors of the AD671 need to be externally adjusted to zero. This is accomplished by trimming the voltage at BPO/UPO (Pin 21) and REFIN (Pin 19). In those applications the AD588, a high precision pin programmable voltage reference, is an ideal choice. The AD588 includes a reference cell and three additional amplifiers which can be configured to provide offset and gain trims for the AD671. The circuit in Figure 7 is recommended for calibrating offset and gain errors of the AD671 when configured in the 0 V to +10 V input range.
+ 5V 10F -5V 10F + 5V 10F
0.1F
0.1F
0.1F
0 TO +10V +15V 39k R1 100
23 VCC 20 AIN
24 VEE
17 VLOGIC BIT1 12 BIT12 1
22 ACOM 150pF 3 1 14 1F 10F 10k 0.1F 0.1F 21 BPO/UPO 18 DCOM 50 19 REF IN
ENCODE 16 DAV 15 OTR 14 MSB 13
0.1F
0.1F
0.1F
1F
7
6
4
23 VCC + 20 AIN VIN 5V - AGP* DGP* + 5V REF 22 ACOM 18 DCOM
24 VEE
17 VLOGIC
AD588
BIT1 12 BIT12 1
5 9 10 8 12 11 R2 100k 5k 13
15 2 16
AD671
150 10F +15 -15
ENCODE 16 DAV 15 OTR 14 MSB 13
50
19 REF IN 21 BPO/UPO
Figure 7. Unipolar (0 V to +10 V) Calibration
AD671
*GROUND PLANE RECOMMENDED
Figure 6. AD671 Grounding and Decoupling
The AD671 is intended to have a nominal 1/2 LSB offset so that the exact analog input for a given code will be in the middle of that code (halfway between the transitions to the codes above it and below it). Thus, the first transition ( from 0000 0000 0000 to 0000 0000 0001) will occur for an input level of +1/2 LSB (1.22 mV for 10 V range). If the offset trim resistor R2 is used, -9-
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AD671
it should be trimmed as above, although a different offset can be set for a particular system requirement. This circuit will give approximately 50 mV of offset trim range. The gain trim is done by applying a signal 1 1/2 LSBs below the nominal full scale (9.9963 for a 10 V range). Trim R1 to give the last transition (1111 1111 1110 to 11111111 1111).
UNIPOLAR (0 V TO +5 V) CALIBRATION
Bipolar calibration is similar to unipolar calibration. First, a signal 1/2 LSB above negative full scale (-4.9988 V) is applied and R1 is trimmed to give the first transition (0000 0000 0000 to 0000 0000 0001). Then a signal 1 1/2 LSB below positive full scale (+4.9963) is applied, and R2 is trimmed to give the last transition (1111 1111 1110 to 1111 1111 1111).
OUTPUT LATCHES
The connections for the 0 V to +5 V input range calibration is shown in Figure 8. The AD586, a +5 V precision voltage reference, is an excellent choice for this mode of operation because of its performance, stability and optional fine trim. The AD845 (16 MHz, low power, low cost op amp) is used to maintain the +5 volts under the dynamically changing load conditions of the reference input.
+15V 2 0 TO +5V 3 4 -15V 0.1F +15V +15V 2 +VIN 2 VOUT 6 3 7 AD845 4 MSB 13 10k -15V 0.1F 0.1F 6 18 DCOM 19 REFIN DAV 15 OTR 14 7 AD845 1 8 1k 390 22 ACOM ENCODE 16 0.1F 6 23 VCC 20 AIN 24 17 VEE VLOGIC BIT1 12 BIT12 1
Figure 10 shows the AD671 connected to the 74HC574 Octal D-type edge triggered latches with 3-state outputs. The latch can drive highly capacitive loads (i.e., bus lines, I/O ports) while maintaining the data signal integrity. The maximum set-up and hold times of the 574 type latch must be less than 20 ns (tDD and tSS minimum). To satisfy the requirements of the 574 type latch the recommended logic families are HC, S, AS, ALS, F or BCT. New data from the AD671 is latched on the rising edge of the DAV (Pin 24) output pulse. Previous data can be latched by inverting the DAV output with a 7404 type inverter. See Figures 20, 21 and 22 for PCB layout recommendations.
74HC574
BIT1 BIT2 BIT3 BIT4 BIT5 BIT6 BIT7 BIT8 DAV 1D 2D 3D 4D 5D U6 6D 7D 8D CLK 1Q 2Q 3Q 4Q 5Q 6Q 7Q 8Q OC DATA BUS
21 BPO/UPO +15V
TRIM 5 8 NOISE REDUCTION 1F
AD671
AD586
4 GND
Figure 8. Unipolar (0 V to +5 V) Calibration
BIT9 BIT10 BIT11 BIT12
74HC574
1D 2D 3D 4D 5D U5 6D 7D 8D CLK 1Q 2Q 3Q 4Q 5Q 6Q 7Q 8Q OC
The AD671 offset error must be trimmed within the analog input path, either directly in front of the AD671 or within the signal conditioning chain, eliminating offset errors induced by the signal conditioning circuitry. Figure 8 shows an example of how the offset error can be trimmed in front of the AD671. The AD586 is configured in the optional fine trim mode to provide +6%/-2% (+240 LSBs/-80 LSBs) of gain trim. The procedure for trimming the offset and gain errors is similar to that used for the unipolar 10 V range with the analog input values set to onehalf the 10 V range values.
BIPOLAR ( 5 V) CALIBRATION
AD671
3-STATE CONTROL
Figure 10. AD671 to Output Latches
OUT OF RANGE
The connections for the bipolar input range is shown in Figure 9. The AD588 is configured to provide dual +5 V outputs. Providing a +5 V reference voltage for the AD671 gain trim and the +5 V BPO/UPO input for the bipolar offset trim.
5V +15V 39k 150pF 7 6 4 3 1 14 150pF 10F R2 100 10F 0.1F 0.1F 21 BPO/UPO MSB 13 50 19 REF IN OTR 14 6.2k R1 100 22 ACOM 18 DCOM 23 VCC 20 AIN 24 VEE 17 VLOGIC BIT1 12 BIT12 1 ENCODE 16 DAV 15
An Out of Range condition exists when the analog input voltage is beyond the input range (0 V to +5 V, 0 V to +10 V, 5 V) of the converter. OTR (Pin 14) is set low when the analog input voltage is within the analog input range. OTR is set HIGH and will remain HIGH when the analog input voltage exceeds the input range by typically 1/2 LSB (OTR transition is tested to 6 LSBs of accuracy) from the center of the full-scale output codes. OTR will remain HIGH until the analog input is within the input range and another conversion is completed. By logical ANDing OTR with the MSB and its complement overrange high or underrange low conditions can be detected. Table II is a truth table for the over/under range circuit in Figure 11. Systems requiring programmable gain conditioning prior to the AD671 can immediately detect an out of range condition, thus eliminating gain selection iterations.
Table II. Out of Range Truth Table
1F
OTR 0 0 1 1 -10-
MSB 0 1 0 1
Analog Input Is In Range In Range Underrange Overrange REV. B
AD588
15 2 16 50 +15 -15
AD671
5
9
10
8
12
11
13
Figure 9. Bipolar (5 V) Calibration
AD671
MSB OTR OVER = "1"
UNDER = "1" MSB
Figure 11. Overrange or Underrange Logic
OUTPUT DATA FORMAT
The AD671 provides both MSB and MSB outputs, delivering data in positive true straight binary for unipolar input ranges and positive true offset binary or twos complement for bipolar
input ranges. Straight binary coding is used for systems that accept positive-only signals. If straight binary coding is used with bipolar input signals a 0 V input would result in a binary output of 2048. The application software would have to subtract 2048 to determine the true input voltage. Most processors typically perform math on signed integers and assume data is in that format. Twos complement format minimizes software overhead which is especially important in high speed data transfers, such as a DMA operation. The CPU is not bogged down performing data conversion steps, hence increasing the total system throughput.
Table III. Output Data Format
Input Range 0 to +5 V
Coding Straight Binary
Analog Input1 -0.00061 V 0V +5 V >+5.00061 V -0.00122 V 0V +10 V +10.00122 V -5.00122 V -5 V 0V +4.99756 V +4.99878 V -5.00122 V -5 V 0V +4.99756 V +4.99878 V
Digital Output 0000 0000 0000 0000 0000 0000 1111 1111 1111 1111 1111 1111 0000 0000 0000 0000 0000 0000 1111 1111 1111 1111 1111 1111 0000 0000 0000 0000 0000 0000 1000 0000 0000 1111 1111 1111 1111 1111 1111 1000 0000 0000 1000 0000 0000 0000 0000 0000 0111 1111 1111 0111 1111 1111
OTR2 1 0 0 1 1 0 0 1 1 0 0 0 1 1 0 0 0 1
0 to +10 V
Straight Binary
-5 V to +5 V
Offset Binary
-5 V to +5 V
2s Complement (Using MSB)
NOTES 1 Voltages listed are with offset and gain errors adjusted to zero. 2 Typical performance.
ILOGIC vs. CONVERSION RATE
Figure 12 shows the typical logic supply current vs. conversion rate for various capacitive loads on the digital outputs.
6.5 6.0 5.5 5.0 4.5 4.0
mA
CL = 50pF
CL = 30pF 3.5 3.0 2.5 2.0 1.5 1.0 0.5 1k 10k 100k CONVERSION RATE - Hz 1M 10M CL = 0pF
Figure 12. ILOGIC vs. Conversion Rate for Various Capacitive Loads on the Digital Outputs
REV. B
-11-
AD671
HIGH PERFORMANCE SAMPLE-AND-HOLD AMPLIFIER (SHA) CROSS COUPLED LATCH
In order to take full advantage of the AD671's high speed capabilities, a sample-and-hold amplifier (SHA) with fast acquisition capabilities and rigid accuracy requirements is essential. One possibility is a hybrid SHA such as the HTC-0300A, but often a cost effective alternative like the one shown in Figure 13 may be a better solution. This discrete SHA requires very few components and is able to acquire signals to 0.01% accuracy in less than 350 nanoseconds. Combined with the AD671, signals with bandwidths up to 500 kHz can be converted with 12-bit accuracy.
R9 R7 1k +15V VIN (5Vp-p) R6 4 2k C24 R8 250 4 IN1 5 IN2 13 IN3 R11 250 -15V R10 10k D1 1N4148 2 1k C28 20pF +15V C26 SD5001 OUT1 1 OUT2 8 2 7 U9 AD845 4 3 0.1F 6 C27
As noted in the Theory of Operation, the ENCODE pulse is specified to operate within a window of time. The circuit in Figure 14 can be used to generate a valid ENCODE pulse if a clock pulse width of greater than 30 ns is available.
1/4 7402
tw
AD671
1/4 7402
ENCODE
1/4 7402
DAV
Figure 14. Cross Coupled Latch
TIMING DESCRIPTION
11 U8 10 AD841 6 C25 5 0.1F -15V
0.1F
U10
OUT3 16 OUT4 9 12 IN4 G1 G2 G3 G4 3 6 14 11
C29 20pF R13 1k
-15V 0.1F C34 5pF R14 226
S/H VR2 100k S/H PEDESTAL ADJ
Figure 15 shows the timing requirements for the discrete SHA. The complementary S/H inputs are HCMOS-compatible although larger gate voltages will improve performance by lowering the on resistances of the DMOS switches. It should be noted that a conversion is started before the SHA has settled to 0.01% accuracy. The discrete SHA takes advantage of the fact that the AD671 does not require a 12-bit accurate input until it is 150 ns into its conversion cycle. See Figures 21, 22 and 23 for PCB layout recommendations.
t SAMPLE = 1s
Figure 13. Discrete High Speed Sample-and-Hold Amplifier
CIRCUIT DESCRIPTION
The discrete SHA shown in Figure 13 is a closed-loop, noninverting architecture which accepts 5 V p-p inputs. The overall gain of the SHA is +2 in order to accommodate the 10 V input span of the AD671. The AD841, with a 0.01% settling time of 110 ns, is the suggested input buffer to the SHA. The circuit also employs a SD5001 which contains four ultrahigh speed DMOS switches (Q1-Q4). The high CMRR, low input offset current, and fast settling time of the AD845 op amp are all critical features necessary for optimal performance of the discrete SHA. In sample mode, Q1 and Q3 of the SD5001 are closed (Q2 and Q4 are open). C28 is charged to the input voltage level at a rate primarily determined by the time constant, R9 * C28. Simultaneously, C29 is connected to ground through a 250 ohm resistor. If C28 is equal to C29, charge injection from Q1 will be approximately equal to charge injection from Q3 based on the symmetry of the circuit and the inherent matching of the switch capacitances. The resultant pedestal errors appear as a commonmode signal to the AD845. VR2, R13, R14, and C34 may be included if further reduction of pedestal error is required. In hold mode, Q2 and Q4 are closed (Q1 and Q3 are open) to reduce feedthrough. The input signal is attenuated -78 dB relative to the input signal at frequencies up to 500 kHz. The AD845 buffers the voltage on C28 and also provides the wideband, low-impedance output necessary to drive the input of the AD671. Droop, which occurs as a result of leakage currents, will appear on C28 and will similarly appear on C29. Like pedestal errors, droop appears as a common-mode signal to the AD845 and is greatly reduced by the differential nature of the circuit. Voltage droop is typically 5 V/s.
ENCODE
t CONVERSION = 500ns
DAV
t ACQUIRE
350ns S/H
t SETTLE
350ns
Figure 15. AD671 to Discrete SHA Timing Diagram
DYNAMIC PERFORMANCE
In most sampling applications the dynamic performance of the system is limited by the performance of the SHA. The SHA's dynamic performance can be selected to meet the system sampling requirements. Figures 16 and 17 are typical FFT plots using the discrete SHA in Figure 13.
Figure 16. Typical FFT Plot of AD671 and Discrete SHA FIN = 100 kHz
-12-
REV. B
AD671
DYNAMIC CHARACTERISTICS
(@ +25C, tested using the discrete SHA in Figure 15 with V CC = +5 V, VLOGIC = +5 V, VEE = -5 V, f SAMPLE = 1 MSPS)1
Model Effective Number of Bits (ENOB) FIN = 100 kHz FIN = 490 kHz
AD671JD-500 Typ Units 11.3 11.2 Bits Bits dB dB dB dB dB dB
Figure 17. Typical FFT Plot of AD671 and Discrete SHA FIN = 500 kHz
Signal-to-Noise and Distortion (S/N+D) Ratio FIN = 100 kHz 70 FIN = 490 kHz 68 Total Harmonic Distortion (THD) FIN = 100 kHz FIN = 490 kHz Peak Spurious (dc to 490 kHz) Peak Harmonic Component (dc to 490 kHz) -80 -75 -79 -76
MULTICHANNNEL DATA ACQUISITION SYSTEM
The AD684, a quad high speed sample-and-hold amplifier is ideally suited for multichannel data acquisition applications. Figure 18 shows a typical data acquisition circuit using the AD684 (SHA), ADG201HS (Multiplexer), AD588 (Reference) and the AD671. The AD684 is configured to simultaneously sample four analog inputs. Each held analog input voltage can be selected by the multiplexer and buffered by the AD841. The AD671 is connected in the bipolar input range ( 5 V).
NOTE 1 fIN amplitude = -0.2 dB @ 100 kHz and -0.9 dB @ 490 kHz, bipolar mode unless otherwise indicated. See Definition of Specifications for additional information.
Figure 18. Data Acquisition System Using the AD684 and the AD671
REV. B
-13-
AD671
AD671 TO ADSP-2100A INTERFACE
Figure 19 demonstrates the AD671 to ADSP-2100A interface. The 2100A with a clock frequency of 12.5 MHz can execute an instruction in one 80 ns cycle. The AD671 is configured to perform continuous time sampling. The DAV output of the AD671 is asserted at the end of each conversion. DAV can be used to latch the conversion result into the two 574 octal D-latches. The falling edge of the sampling clock is used to generate an interrupt (IRQ3) for the processor. Upon interrupt, the ADSP2100A starts a data memory read by providing an address on the DMA bus. The decoded address generates OE for the latches and the processor reads their output over the DMA bus. The conversion result is read within a single processor cycle.
AD671 TO ADSP-2101/ADSP-2102 INTERFACE
DMRD DMA0:13 ADDRESS BUS
DAV OE 8
574
Q0:7 8 D0:7 OE
AD671
DECODE
ADSP-2100A
16 DMA0:15 DMACK +5V DATA BUS 8 Q0:7 D0:7 IRQ3 SAMPLING CLOCK BIT1:12 4 4 ENCODE
574
D0:3
Figure 19. AD671 to ADSP-2100A Interface
Figure 20 is identical to the 2100A interface except the sampling clock is used to generate an interrupt (IRQ2) for the processor. Upon interrupt the ADSP-2101A starts a data memory read by providing an address on the Address (A) bus. The decode address generates OE for the D-latches and the processor reads their output over the Data (D) bus. Reading the conversion result is thus completed within a single processor cycle.
RD A0:13 ADDRESS BUS
DAV OE 8
574
Q0:7 8 D0:7 OE
AD671
DECODE
ADSP-2101
16 D0:15 DATA BUS 8 Q0:7 D0:7 IRQ2 SAMPLING CLOCK BIT1:12
574
D0:3 4
4
ENCODE
Figure 20. AD671 to ADSP-2101/ADSP-2102 Interface
Figure 21. PCB Silkscreen and Component Placement Diagram for Figures 5, 10 and 13
-14-
REV. B
AD671
Figure 22. PCB Solder Side Layout for Figures 5, 10 and 13
Figure 23. PCB Component Side Layout for Figures 5, 10 and 13
REV. B
-15-
AD671
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
24-Pin Plastic DIP (Suffix N)
24-Pin Ceramic DIP (Suffix D)
PIN 1 1 1.200 (30.48 0.012 0.31)
0.295 (7.49
0.01 0.26) 0.300 (7.49 0.010 0.25)
SEATING PLANE 0.175 (4.45) 0.018 0.002 (0.46 0.05) TYP 0.100 (2.54 0.005 0.13) 0.05 (1.27) TYP
0.085 (2.16
0.009 0.23)
0.010
+ 0.002
TOLL NON ACCUM NOTES 1. LEAD NO. 1 IDENTIFIED BY DOT OR NOTCH. 2. CERAMIC DIP LEADS WILL BE EITHER GOLD OR TIN PLATED IN ACCORDANCE WITH MIL-M-385 TO REQUIREMENTS.
-16-
REV. B
PRINTED IN U.S.A.
1.100 (27.94
0.005 0.13)
(
-0.001 + 0.05 0.025 -0.03
)
C1426a-10-9/91

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